1、咸寧學院本科畢業論文：外文翻譯電子與信息工程學院 本科畢業論文(設計)外 文 文 獻 翻 譯譯文題目：single-stage single-switch powerfacter correction ac/dc converter學生姓名： 專 業： 指導教師： 2010年 3 月原文:single-stage single-switch power factor correctionac/dc converterabstracta single-stage single-switch power factor correction ac/dc converter is proposed i

2、n which the power factor correction (pfc) inductor is connedted to a tap on the primary winding ofthe dc/dc flyback converter stage; there is direct energy transfer of a part of the input energy tothe output and thedc-bus voltage feedback. the additional discharge path in the pfc inductorand dc-bus

3、voltage feedback effectively suppresses the dc-bus voltage and increases the overallefficiency. experimental results for a 60w converter at a constant switching frequency of 70 khzare obtained to show the performance of the proposed converter. it is shown that the voltageacross the dc-bus capacitor

4、can be held below 405v even though the converter operates in a widerange of input voltages (90v-265vac) and the measured input current harmonics satisfy the iec。1 introductionmany pfc ac/dc converters have been presented inrecent years. pfc techniques can be divided into twocategories: a single-stag

5、e approach and a two-stageapproach. the two-stage approach is the most commonlyused approach. conventional two-stage pfc convertersinvolve the use of cascading two power-processing stages,responsible for power factor correction and output voltageregualtion. a pfc converter is adopted at the front-en

6、d toforce the line current tracking the line voltage and anotherconventional dc/dc converter is cascaded after the pfcstage to obtain the desired tightly regulated output voltage.this approach can obtain very good performance, such ashigh power factor, and low voltage stress. however, due tothe use

7、of two power-processing stages, conversionefficiency is reduced and an additional pfc stage addscomponents and complexity. consequently the overall costincreases. the two-stage approach has disadvantages oflowpower density, control complexity, and high cost.to reduce the overall size and cost, a num

8、ber of siglestagepfc converters have been developed in the literature witches so that the pfc switch anditscontroller can be eliminated. thesingle-stage approach is specially attractive in low-cost and low-power applicationsdue to its simple power stage and control circuit. however,it still has seve

9、ral drawbacks such as high current stress inpower switch and high dc-bus voltage stress. its majordrawback is a high voltage stress on the dc-bus capacitor.many single-stage pfc ac/dc converters suffer from highdc-bus voltage stress at light load and high line, whichmakes these converters impractica

10、l. a high dc-busvoltage means high component rating, high cost and lowvoltage stress. experimental results for a 60w converter ata constant switching frequency of 70 khz are obtained toshow the performance of the proposed converter. it is hown that the voltage across the dc-bus capacitor can beheld

11、below 404v even though the converter operates in awide range of input voltages (90v-265vac) and themeasured input current harmonics satisfy the iec 61000-3-2class d requirements.2 analysis of proposed converter figure 1 shows the equivalent circuits of the proposedconverter. the secondary winding nb

12、 is added in the pfc boost inductor. the first transformer t1 in dc/dc part can be operated either in ccm or dcm depending on the load conditions as in the conventional flyback converter. for simplification, dc/dc part is assumed to operate in ccm for entire line period. the second transformer t2 is

13、 operated in dcm. according to the operation of t2, when the dc-bus voltage feedback value v1(=np2vab, np2=n2/np) is higher than the rectified line input voltage vi, the converter operation enters region a. in region a only t1 is operating and db is reverse-biased during the on-time period s. when v

14、i is higher than v1 and lower than v2 the operation converter enters region b. in this region t1 and t2 work like flyback transformers; v2 is determined by vd+(n1-na)vo where n1=n1/ns and na=na/nb.when vi is higher than v2 the converter operation enters region c. in this region t1 works like a flyba

15、ck transformer and t2 works like a boost inductor. since the voltage vi is vm sin ot in the first quarter of line period, the boundary times tx and ty for three modes are given byfig. 1 equivalent circuit2.1 region a operationin region a the input voltage vi is lower than the dc-bus voltage feedback

16、 value vl=(n2/np)vd. only the dc/dc part operates. it delivers power from the dc-bus capacitor cd to the load rl through t1. at to the switch s is turned on. since the dc-bus voltage vd is applied across the magnetising inductance lm1, the magnetising current ilm1 increases linearly from its lowpeak

17、 value ima,l as follows:at t1 the switch s is turned off and the output diode d1 is on. since -npvo is applied across the magnetisinginductance lm1, the current decreases linearly from its highpeak value ima,h as follows:the diode current id1 is given byfrom (3) and (4) the voltage gain is determine

18、d as follows:since the dc/dc part operates in ccm, the duty cycle d does not change with the load variation. from (6) the turn ration np can be determined bythe duty cycle d can be obtained asthe duty cycle d is also effective in regions b and c. the power p1 delivered by the transformer t1 is deter

19、mined assince the power p1 delivered by the transformer t1 should be equal to the output power po (=vo2 /rl), the following relation is obtained:consequently the dc-bus capacitor provides the whole power to the load. from (3), (6) and (10), the high and low peak values of ilm1 are determined by2.2 r

20、egion b operationwhen the voltage vi is higher than v1 and lower than v2, the converter operates in region b. pfc cell operates as dcm flyback converter. at t0 the switch s is turned on. since the dc-bus voltage vd is applied across the magnetising inductance lm1, the magnetising current ilm1 increa

21、ses linearly from its low peak value imb,las follows:the magnetising current ilm2 increases linearly from zero as follows:sincethe dc-bus capacitor current icd, the current i2, and the switch current is are given by from (15) it can be seen that the magnetising current ilm1 is supplied by the magnet

22、ising current ilm2 and the discharging current icd of the dc-bus capacitor cd. from (17) the switch current is is also composed of the two components. the condution loss can be reduced by selecting smaller np1. at t1 the switch s is turned off and the output diodes d1 and d2 are on. since npvo is ap

23、plied across the magnetising inductance lm1, the current ilm1decreases linearly from its high peak value imb,h asfollows:since the diode d2 is on, _navo is applied across the magnetising inductance lm2 and the current ilm2 decreases linearly from its peak value im,p as follows: the peak value im,p i

24、s given byfrom (20)at t2 the current ilm2 arrives at zero and the diode d2 isturned off. the power p1 delivered by the transformer t1 is given bythe power p2 delivered by the transformer t2 is determined bysince the sum of the power p1 delivered by the transformer t1 and the power p2 delivered by th

25、e transformer t2 should be equal to the output power po, the following relation isobtained:from (6), (13) and (24), the high and low peak values of ilm1 are determined by2.3 region c operationwhen the voltage vi is higher than v2 the converter operates in region c. the pfc cell operates as a dcm boo

26、st converter and the diode d2 is off in this region. at t0 the switch s is turned on, ilm1 increases linearly asfollows:ilm2 increases linearly as in region b. equations (9)(11) are also effective in this region. the magnetising current ilm1 is supplied by the magnetising current ilm2 and the discha

27、rgingcurrent icd of the dc-bus capacitor cd. when the current ilm2 is high enough that np2ilm2 exceeds ilm1, the dc-bus capacitor can be in charging mode during the ontime interval of s. this charging mode may occur near the line peak voltage as follows:at t1 the switch s is turned off and the outpu

28、t diodes d1.since _npvo is applied across the magnetising inductance lm1, the current ilm1 decreases linearly from its high peak value imc,h as follows:since the diode db is on, the voltage across the magnetising inductance lm2 is _vd_n1vo+vi and the current ilm2 decreases linearly from its peak val

29、ue im,p as follows:the peak value im,p is given byfrom (31)sincethe current i1, the dc-bus capacitor current icd, and the diode current id1 are determine as follows:the current id1 is composed of two components which are from lm1 and lm2. therefore, there is direct power transfer from the line input

30、 to the load during the switch off-time. as a result the overall efficiency can be improved. at t2, the current ilm2 arrives at zero and the diode db is turned off. the current id1 is npilm1. since the output power should be equal to the sum of the power from lm1 and the power from lm2, the followin

31、g relation isobtained:from (6), (13) and (24), the high and low peak values of ilm1 are determined by3 concluding remarksa single-stage single-switch power factor correction ac/dcconverter has been proposed. experimental results for a60wconverter at a constant switching frequency of 70 khz have been

32、 given to show the performance of the proposed converter. experimental results have shown that the voltageacross the dc-bus capacitor can be held below 404v even though the converter operates in a wide range of inputvoltages (90b265 v). as a result, commercially availableelectrolytic capacitors, can

33、 be used. the ccm dc/dcstage, direct energy transfer and the reduced conduction lossin the power switch can increase the efficiency and itsmaximum value is 91.1%. the converter meets en/iec61000-3-2 class d requirements.譯文:單級單刀功率因數校正器 ac / dc變換器 摘要單級單刀開關ac / dc功率因數校正器（pfc）是電感連接到初級繞組的直流/直流轉換器；輸入能量的一部

34、分直接轉移成輸出能量和反饋給dc總線。另外在pfc電感放電和dc總線電壓反饋過程中，有效地抑制了直流母線電壓，提高了整體效率，在一個功率60w、恒定開關頻率70千赫轉換器的實驗中的性能結果表明，電壓在直流總線下電容可以兼容，即使低于405v，轉器工作在帶寬輸入電壓（90-265 vac）的條件下，測量輸入電諧頻各項指標滿足國際電工委員會要求。1簡介近年來，許多功率因數校正ac / dc轉換器相繼問世。 pfc的技術可以分為兩個類別：單級方法和雙級方法。這兩個不同層次的方法是最常使用的方法。傳統的雙級pfc變換器涉及兩個級聯在電路中使用，負責功率因數校正和輸出額定電壓 。pfc轉換器是采用在前端

35、迫使線電流跟蹤線電壓，一種傳統的直流/直流轉換器串聯后的pfc可以獲得現階段所需的穩定的額定輸出電壓。這種方法可以取得很好的成效，如提高功率因數，降低電壓沖擊。然而，由于兩種電源的工作轉換使用效率降低，增加了pfc階段額外的組件和復雜性,因此整體成本增加。這兩個階段的方法具有低功率強度，控制復雜，成本高的缺點。 為了減少整體規模和成本，在一些相關著作中對很多單級pfc的轉換器作了詳細闡述。其主要思想是，pfc和dc / dc有共同的pfc開關，它的開關及其控制器可以被閑置。由于單級方法簡單特別是在低成本和低功耗地方得到了應用。然而，它仍然有幾個缺點 比如在電源開關和高直流母線電壓沖擊下有較高的

36、電流脈動。其主要缺點是對直流總線電容有高電壓損傷。許多pfc和c / dc轉換器受到低負載和高直流限制，高直流總線高電壓組合方法具有高成本和低效率的特點。橫跨直流總線電容的電壓隨著輸入電壓和負載的變化而變化，特別是當在不連續工作的傳導模式（dcm）和直流/直流中的部分 pfc是連續傳導模式（ccm）。dcm pfc和 ccm的直流/直流組合可生成一個高達1000v的直流母線電壓它可以像對在高線和低荷載線路上普遍應用 。然而，在dcm或者ccm兩部分的結合操作模式中沒有直流母線電壓應力的問題。由于dcm的升壓轉換器有其內在的pfc特性， pfc部分更適合在dcm中操作 。如果pfc部分和dc/d

37、c部分都在dcm中操作，則直流總線電容的電壓是獨立的負載。然而，與ccm操作模式相比較，高開關電流需要高電流額定開關并且會降低效率。因此，在低功耗應用中dcm pfc和ccm dc/ dc的結合被認為是理想的組合。為了抑制高直流母線電壓應力，一個變頻調速被很好的使用起來，但是它也存在一些問題，類似于高頻率和電感元件及輸入濾波器的設計上有問題。pfc的附加繞組電感能有效地抑制直流母線電壓并且提供直接能源轉換。然而，當開關打開，無論是升壓電感電流和磁化電流都會流過開關。也就是說，開關電流沖擊是pfc的一部分，它是組成直流/直流的部分，它增加傳導損耗。提出了單級單開關功率因數校正ac / dc轉換器

38、，因為第二繞組中加入pfc升壓電感，輸入功率是直接轉化為輸出功率。此外，在dc/dc級繞組的初級階段通過連接上的pfc電感扼流圈一直到首端，有一部分的輸入能量直接轉化成輸出能量。并且，當開關為on時，在直流/直流變壓器轉化時有一些升壓電感的磁化會使電感充電，它可以減少傳導損失。一般來說，這些直接的能量轉移會提高整體效率和抑制直流總線電壓。一個60w的轉換器的實驗中，恒定開關頻率為70千赫換器的性能結果表明，電壓在直流總線電容可以兼容，即使低于405v，轉換器工作在寬輸入電壓范圍（90b265vac）的條件下，測量輸入電諧頻各項指標滿足國際電工委員會要求。2轉換器的分析建議圖1顯示了轉換器的等效

39、電路圖。二次繞組中添加nb ，是pfc 的升壓電感。一次側變壓器t1在dc / dc部分無論是ccm或dcm，可以根據負載在常規條件下反饋給轉換器。為了簡化，直流/直流部分用于ccm總線路的操作中。變壓器t2的在dcm中。根據t2的運作，有三個操作，如圖3所示，當直流總線v1的電壓反饋值 (=np2vd, np2=n2/np)超過輸入電壓的整流線的電壓，該轉換器就會在區域a運行。在a區只有t1是正在運行，db是在一段時間s上反向偏置。vi高于v1和低于v2的時候，此時轉換操作進入b區。在這個區域，t1和t2工作起來就像一個回掃轉換器; v2是由vd+(n1-na)vo決定。其中n1=n1/ns

40、 和na=na/nb。當viv2時該轉換器操作進入c區。在這個區域，t1就像一個回掃變壓器在工作并且t2就像一個升壓電感。由于vi電壓是正弦曲線最大值的四分之一，是介于tx和ty的，如下所示 ： 圖1轉換器的等效電路圖 2.1 a區操作在區域a，輸入電壓vi低于直流總線v1的電壓反饋值，vl=(n2/np)vd.，只有直流/直流部分運作。它提供的直流電源通過t1使總線電容從cd的負荷傳給rl。在t0時刻開關s是打開的。由于直流母線電壓vd適用于整個磁化電感lm1中，磁化電流ilm1從最低的ima，l的峰值線性增加如下：磁化電感l，從最在t1時刻開關s是關閉的，輸出二極管d1處于工作狀態。由于npvo是適用于整個高的ima，h的峰值線性減少如下：二極管電流id1是：在（3）及（4）中電壓v0 如下： 由于直流/直流部分在ccm中運作，環流d不會改變負載的變化。從（6）看出np可確定為：環流d的表達式為：環流d 在b區和c區也是